This invention relates to the art of electric current measurement, and especially to the art of utilizing ordinary current transformers to measure a-c (alternating-current) current, d-c (direct-current) current, and a-c current having a d-c current component. The invention disclosed herein utilizes a common electronic control circuit to provide magnetic control for multiple current transformers, thereby providing a way to accurately measure multiple electric currents for lower cost than competing technologies.
The present invention is a development of the inventions disclosed in U.S. Pat. No. 6,160,697 to Edel, issued Dec. 12, 2000; U.S. Pat. No. 6,522,517 to Edel, issued Feb. 18, 2003; and U.S. Pat. No. 7,242,157 to Edel, issued Jul. 10, 2007. These patents will be referenced herein as “the 697 patent”, “the 517 patent”, and “the 157 patent” respectively.
U.S. Pat. No. 6,522,517 to Edel, issued Feb. 18, 2003, (the 517 patent) is hereby incorporated by reference, in its entirety, into this disclosure.
The 697 patent and the 517 patent disclose how a varying voltage, produced by a “controllable voltage device,” may be applied to a conductive winding that magnetically interacts with a magnetic body. The varying voltage controls the voltage induced in the winding in such a way that the integral over time of the induced voltage correlates to desired changes of the induction of the magnetic body. (As used in this disclosure, the term “induction level” is synonymous with the terms “magnetization” and “magnetic flux density”). The induction level may be controlled in several ways:    (a) The induction level of a magnetic body may be caused to transition from a known induction level to a preferred induction level. (A preferred induction level of zero may be chosen to demagnetize a magnetic body).    (b) When the induction level is not known, a preferred induction level may be established by changing the induction level of the magnetic body from an unknown induction level to a known induction level and then to the preferred induction level.    (c) A preferred induction level may be maintained by causing the induced voltage across the winding to have an average value near zero (or by causing the integral of induced voltage to not exceed a predetermined value).    (d) A preferred induction level may be more strictly maintained by causing the induced voltage across the winding to continuously be near zero volts, thereby reducing the amount that the induction level fluctuates.    (e) The induction level may be made to vary with time in a preferred manner, including matching a control signal that is proportional to a reference induction level.
The varying voltage may be generated directly by an active voltage source, or the varying voltage may be generated indirectly, such as by current transformer secondary current flowing through an adjustable impedance. The key elements are a magnetic body, a conductive winding that magnetically interacts with the magnetic body, and a means of causing the induced voltage to have the appropriate waveform and magnitude.
A conductive winding that is utilized for controlling induction level may be a permanent winding that is also used for other purposes, or it may be a dedicated winding (permanent or temporary) provided solely for the purpose of controlling induction level.
These principles are most readily applied to magnetic bodies that are configured to have a relatively uniform magnetic path, such as the magnetic cores of current transformers. Using the principles of the 517 patent, the accuracy of a current transformer may be improved in three ways:    (a) By demagnetizing a current transformer, inaccuracies associated with core magnetization are removed. A demagnetized current transformer can accurately measure d-c current and a-c current that has a d-c component.    (b) By keeping the integral over time of induced voltage near zero, a current transformer is better able to measure unsymmetrical currents without quickly transitioning to saturation.    (c) By reducing the amount that the magnetic flux in the core fluctuates, inaccuracies associated with magnetizing current may be greatly reduced and the accuracy of a current transformer may be greatly improved.
Some embodiments of the present invention utilize voltage pulses to control the magnetization of a current transformer magnetic core, rather than a continuously adjustable voltage (as discussed in the 517 patent). The 157 patent discusses this type of switched-voltage magnetic control. Since the 157 patent incorporates the 517 patent by reference, the 157 patent cannot be incorporated herein by reference (per patent office rules). So, for completeness, it is necessary to include herein some of the information contained in the 157 patent.
FIGS. 1 to 9 provide prior-art background information. FIGS. 1 and 2 illustrate ordinary current transformers applied without electronic assist. FIGS. 3 to 9 illustrate ordinary current transformers applied with electronic assist in accordance with the 517 patent and the 157 patent. FIGS. 3 to 9 are similar to FIGS. 1 to 7 of the 157 patent. FIGS. 3 and 4 are also similar to FIGS. 1 and 21 of the 517 patent.
FIG. 1 illustrates a simple prior-art current transformer circuit for measuring a-c current. A current transformer CT comprises a magnetic core 12 and a secondary winding 13.
An electric power system conductor 10 has an insulating covering 11. Power system conductor 10 functions as a primary winding of current transformer CT, with only one turn, with a primary electric current J1 flowing in it. Though shown with one end disconnected, power system conductor 10 is normally connected as part of an electric power system.
For clarity, current transformer CT comprises a magnetic core 12 and a secondary winding 13 magnetically coupled to magnetic core 12; a primary (or “first”) electric current J1 flows in conductor 10 which is configured as a primary winding of current transformer CT; a secondary electric current J2 flows in secondary winding 13; secondary electric current J2 is generally proportionally smaller than primary electric current J1 by a turns ratio of current transformer CT. (Generally speaking, if a current transformer secondary current is known, then the primary current is calculable as the secondary current multiplied by the turns ratio of the current transformer).
As shown in FIG. 1, secondary winding 13 and resistor R1 are connected in series, so that secondary electric current J2 flows through each of them. A ground connection 14 provides a reference for voltage measurements.
Resistor R1 is a current-sensing resistor with low resistance, and may be thought of as one possible embodiment of a “current-sensing means.” Voltage V3, across resistor R1, is proportional to secondary electric current J2, and may be considered to be an information signal that contains information about secondary electric current J2. Resistor R1 functions as a current-sensing means for sensing secondary electric current J2 and provides an information signal containing information about secondary electric current J2.
Winding 13 is shown with ten turns around magnetic core 12. The actual number of turns may vary widely depending on the application. Magnetic core 12 is shown as a solid-core toroid, though wide variation in magnetic core configurations is possible and the illustration is not intended to limit the breadth of application of the invention. Of particular note is the possible use of split-core current transformers, which can be installed around existing conductors without having to thread one end of the primary conductor through the window opening of the current transformer.
A measurement circuit 15 comprises an analog-to-digital circuit 22 and a microcontroller circuit 23. Analog-to-digital converter circuit 22 has a high-impedance voltage-sensing input (connected to conductor 4) to enable sensing of voltage V3 relative to the potential of conductor 5. Analog-to-digital converter circuit 22 receives the information signal (voltage V3) via conductors 4 and 5, and converts the signal to a digital information signal for use by microcontroller 23. Analog-to-digital converter circuit 22 and microcontroller 23 communicate via an interface shown as four conductors, and this interface may vary considerably depending on the particular design.
Microcontroller 23 processes the digital signal, and typically communicates information about the measured current to other equipment or a visual display via multiple conductors 16. Microcontroller 23 may be part of a larger electric power monitor which also measures voltage, in which case microcontroller 23 may utilize the information about the electric current to calculate electric power or electric energy parameters.
FIG. 2 illustrates prior art non-contact a-c current measurement of multiple a-c currents with no electronic assist. FIG. 2 is similar to FIG. 1, except that FIG. 2 shows multiple current transformers (each labeled CT) for measuring multiple primary electric currents (each labeled J1) flowing in multiple power system conductors (each labeled 10). Many components are the same as previously shown, and these components function in the manner previously described. Multiple current-sensing resistors (each labeled R1) are now shown, one for each current transformer. Analog-to-digital converter circuit 22 now has multiple input channels, one for each current-sensing resistor. Analog-to-digital converter circuit 22 may be configured to sample each input channel continuously, or it may be configured to sample only one input channel at once. The number of current transformers and current-sensing resistors connected to a single measurement circuit 15 may vary widely, depending on the application.
FIG. 3 illustrates the general configuration of prior art current measurement utilizing electronic assist, as described in the 517 patent. A controllable voltage device 6 provides an output voltage V1 for magnetic control of current transformer CT. This configuration makes it possible to utilize ordinary current transformers to measure d-c currents and a-c currents having a d-c current component (d-c currents having one or more a-c current components are considered to be the same as a-c currents having a d-c component). Improved accuracy in the measurement of a-c currents is also possible.
Controllable voltage device 6 is connected in series with secondary winding 13 and current-sensing resistor R1, so that secondary electric current J2 flows through each of them. A control circuit 3 now is responsible for both receiving and processing the information signal (voltage V3) and controlling controllable voltage device 6. The dashed lines 60 between control circuit 3 and controllable voltage device 6 indicate control mechanisms by which control circuit 3 controls the connected device (this is the typical meaning of dashed lines connecting from a control circuit throughout this disclosure). The actual control mechanism may vary widely, depending on the particular embodiment.
Control circuit 3 has a high-impedance voltage-sensing input (connected to conductor 4) to enable sensing of voltage V3 relative to the potential of conductor 5. As before, voltage V3 across resistor R1 is proportional to secondary electric current J2. Voltage V2 is the voltage across secondary winding 13, which is the induced voltage generated by changing flux in magnetic body 12 plus any voltage drop associated with current J2 flowing through stray winding impedances. Often, the voltage drop associated with secondary current J2 flowing through stray winding impedances is small compared to the induced voltage and may be ignored in some applications. Voltage V1 is the adjustable output voltage of controllable voltage device 6. Please refer to the 517 patent for additional information.
FIG. 4 illustrates a preferred embodiment of the controllable voltage device of FIG. 3, in accordance with the 517 patent. This configuration provides a continuously adjustable bipolar voltage output (output voltage V1) for magnetic control. Many components are the same as previously shown, and these components function in the manner previously described.
Control circuit 3A has an analog-to-digital converter circuit 22 to sense secondary current J2 (as voltage V3 across current-sensing resistor R1), a microcontroller 23 for data processing and control functions, and a digital-to-analog converter circuit 24. Digital-to-analog converter circuit 24 provides an analog voltage signal on conductor 25 that controls the voltage output of controllable voltage device 6A. Analog-to-digital converter circuit 22 and microcontroller 23 communicate via an interface shown as four conductors, and this interface may vary considerably depending on the particular design. Likewise microcontroller 23 communicates with digital-to-analog converter circuit 24 via an interface shown as four conductors, and this interface may vary considerably depending on the particular design. Alternately, the analog-to-digital converter and digital-to-analog converter may be an integral part of the microcontroller.
Controllable voltage device 6A has a voltage control circuit 1A, and a power supply circuit 2A. Voltage control circuit 1A has an operational amplifier 21, with resistors R7 and R8 configured to set the gain of operational amplifier 21. Power supply circuit 2A provides operating power to operational amplifier 21. A positive d-c voltage is provided at terminal VP, a negative d-c voltage is provided at terminal VN, both relative to a common voltage terminal VG.
Control circuit 3A also requires a power supply circuit, which is not shown. These power supply circuits are well-established in the prior art.
Ground connection 14 provides a common reference potential for the various circuits and power supply. If a particular application requires that winding 13 be directly grounded on one side, then resistor R1 may be relocated and connected in series with the opposite side of winding 13. This complicates the measurement of voltage V3 across resistor R1 somewhat, but prior-art differential voltage measurement methods are adequate and may be used.
Operational amplifier 21 must be able to produce voltage in a circuit with relatively large current driven by a current source (a current transformer acts like a current source). A “power operational amplifier” will usually be required, as discussed in the 517 patent.
The type of control configuration shown in FIG. 4 (utilizing an analog-to-digital converter circuit to sense an input signal, a microcontroller to implement a control function based on the input signal, a digital-to-analog converter to produce an analog control signal, and an operational amplifier to produce a voltage proportional to the analog control signal) is well established in the prior art, so additional configuration details will not be described herein. Microcontroller 23 is configured/programmed to implement the control sequences illustrated and discussed in the 517 patent.
FIG. 5 illustrates the use of a power supply circuit 2B and voltage control circuit 1B as a controllable voltage device 6B, providing a switched-voltage output in accordance with the 157 patent. Many components are the same as previously shown, and these components function in the manner previously described. Power supply circuit 2B provides a single d-c (direct-current) voltage output V4 across two terminals: terminal VP, which has a positive polarity relative to second terminal VN. In the configuration shown, the output of power supply circuit 2B does not have any common connection with the rest of the circuit, so that output voltage V4 can be applied so as to result in a positive or negative polarity at the output of voltage control circuit 1B. That is to say, that aside from connections made by electronic switches 26 and 27, the power supply circuit is isolated from the rest of the circuit.
Voltage control circuit 1B functionally comprises two single-pole triple-throw electronic switches which operate in tandem as controlled by control circuit 3. Electronic switches 26 and 27 operate simultaneously to cause voltage V1 to have one of three values: voltage V1 having positive polarity equal in magnitude to power supply voltage V4, voltage V1 having negative polarity and equal in magnitude to power supply voltage V4, and voltage V1 having a magnitude of approximately zero volts (power supply circuit 2B not connected, and electronic switches 26 and 27 providing a low-impedance path for secondary current J2).
Stated another way, voltage control circuit 1B periodically connects power supply output voltage V4 in series with secondary electric current J2. Voltage control circuit 1B also provides a low-impedance path for secondary current J2 during time periods that power supply circuit 2B is not connected in series with secondary current J2. Voltage control circuit 1B and power supply 2B are configured to provide both positive voltage and negative voltage connected in series with secondary electric current J2, the positive voltage and negative voltage being connected one at a time.
Power supply circuit 2B and voltage control circuit 1B operate together to provide an adjustable output voltage (V1) that is effectively a bipolar pulsed voltage.
Speaking of magnetic control in general, the rate of change of magnetic flux in magnetic core 12 is proportional to the voltage induced in secondary winding 13. Generally speaking, in FIG. 5 control circuit 3 controls the voltage pulses at voltage V1 so that the average value over time of the sum of voltage V1 pulses and of voltage drops associated with secondary electric current J2 flowing through loop impedances is approximately equal to the preferred induced voltage in secondary winding 13, thereby causing the rate of change of magnetic flux to be the preferred value.
When using the term “average value over time,” the time period over which the averaging occurs is significant. In the context of induced voltage calculations within this disclosure, this period of time is preferably equal to the length of time of an integral number of complete voltage pulse cycles (including any off time that is part of a full cycle). A time period spanning only one voltage pulse cycle will generally provide the most accurate control.
For optimal control, control circuit 3 utilizes characteristics of loop impedances through which secondary electric current J2 flows to optimize the control of voltage control circuit 1B for improved control of the magnetic flux of magnetic core 12. Control circuit 3 uses information signal V3 along with known characteristics of loop impedances to calculate the induced voltage generated in winding 13 by flux changes in magnetic core 12. The voltage pulses at voltage V1 are controlled so that the average value of the voltage induced in the winding is the preferred value. For accurate control, the magnitude of voltage V1 should be part of the calculation implemented by control circuit 3, either as a known voltage based on fixed power supply voltage V4 and the states of electronic switches 26 and 27, or voltage V1 may be directly measured by control circuit 3 (measurement of voltage V1 by control circuit 3 would require an additional connection not shown).
Loop impedances include stray impedances associated with winding 13, stray impedances in various connecting wires and electronic switches, and the impedance of current-sensing resistor R1.
For many applications it is desirable to minimize flux variations, in which case control circuit 3 is configured to cause the average value over time of voltage V1 pulses to be approximately equal to, and have opposite polarity as, the sum of voltage drops associated with current J2 flowing through loop impedances, thereby causing the average value of induced voltage to be near zero, thereby minimizing flux changes. This type of control is illustrated by FIGS. 9A to 9E (discussed below).
Stated another way, control circuit 3 receives voltage V3 (which is an information signal containing information about secondary electric current J2) and uses this information for controlling voltage control circuit 1B in such a way as to control changes of magnetic flux in magnetic core 12, thereby reducing the secondary current error, whereby secondary electric current J2 is caused to be more accurately proportional to primary electric current J1. Control circuit 3 operates while current transformer CT is in service.
FIG. 6 illustrates a development of FIG. 5 in which power supply circuit 2A has two voltage outputs relative to a common terminal VG (similar to FIG. 4). Many components are the same as previously shown, and these components function in the manner previously described. A positive d-c voltage is available at terminal VP, and a negative d-c voltage is available at terminal VN. Voltage control circuit 1C is now somewhat simplified (compared to FIG. 5), utilizing three single-pole single-throw electronic switches 28, 29, and 30. The configuration of FIG. 6 is functionally similar to the configuration of FIG. 5, but is easier to realize in practice, since the electronic switches are considerably simpler and the power supply no longer requires an isolated output. Voltage control circuit 1C and power supply circuit 2A together form controllable voltage device 6C.
In the configuration of FIG. 6 voltage control circuit 1C still provides just three values for output voltage V1. With just electronic switch 28 closed, voltage V1 has positive polarity equal to the voltage provided by power supply terminal VP. With just electronic switch 29 closed, voltage V1 has negative polarity equal to the voltage provided by power supply terminal VN. With just electronic switch 30 closed, a low-impedance path is provided for current J2, and voltage V1 is approximately zero volts. Only one switch is normally closed at a time. For most applications, the voltage magnitude of the power supply positive and negative d-c voltage outputs will be fixed (not adjustable) and will be approximately equal. An optional ground connection 14 provides a reference voltage level for the circuit.
FIGS. 7A to 7E illustrate the magnetic operation of an ordinary prior-art current transformer without electronic assist, similar to FIG. 1 and FIG. 2. The waveforms are based on the current transformer operating a-c primary current in sinusoidal steady-state conditions at time T1. Time T2 is a reference mark for waveform comparison. Time T3 marks a change to d-c primary current. Time T4 marks the time that the current transformer magnetic core saturates due to the d-c primary current.
FIG. 7A illustrates one possible primary current waveform for primary current J1. This is the same reference waveform as shown in FIGS. 8A and 9A. The waveform is simply a sinusoidal a-c current for two cycles, then changes to a positive d-c current at time T3.
FIG. 7B simply illustrates that voltage V1 is continuously zero for an ordinary current transformer without electronic assist.
FIG. 7C illustrates the magnetic flux density waveform Y (of magnetic core 12), with the vertical axis scaled linearly from −100% (negative saturation) to +100% (positive saturation). The current transformer burden is assumed to be resistive, so flux waveform Y is 90 degrees out-of-phase with primary current J1 (FIG. 7A) while primary current J1 is sinusoidal. The magnetic core is seen to quickly saturate when primary current J1 changes to a d-c current.
FIG. 7D illustrates the waveform of secondary electric current error J2E. The secondary electric current error is related to a build-up of a magnetic flux in magnetic core 12. For purposes of this disclosure, “secondary electric current error” is the instantaneous difference between the actual secondary electric current (J2) and an ideal secondary electric current calculated as the instantaneous primary electric current (J1) divided by the turns ratio of the current transformer (CT). FIG. 7D shows that secondary current error J2E is relatively large (when compared to FIGS. 8D and 9D with electronic assist). The error becomes 100% (when expressed as a percentage of ideal secondary current) at time T4 when the current transformer magnetic core saturates and secondary current J2 goes to zero magnitude.
FIG. 7E illustrates the waveform of secondary electric current J2. The effects of secondary current error J2E are fairly clear: the sinusoidal waveform is time shifted relative to primary current J1, and the current transformer is unable to maintain a d-c current output. A less-apparent magnitude error is also present during a-c current operation (prior to time T3).
FIGS. 8A to 8E illustrate the magnetic operation of a prior-art current transformer with electronic assist using a continuously adjustable output voltage (this is functionally similar to the configuration of FIG. 4 if control circuit 3A operates to compensate for secondary burden).
FIG. 8A illustrates one possible primary current waveform for primary current J1. This is the same reference waveform as shown in FIGS. 7A and 9A.
FIG. 8B simply illustrates that voltage V1 is continuously controlled to compensate for secondary burden. Since the burden (the sum of all loop impedances) is assumed to be resistive, voltage V1 has a waveform that is virtually the same as secondary current J2 (FIG. 8E), which, due to the high accuracy of the assisted current transformer, is virtually the same waveform as primary current J1 (FIG. 8A).
FIG. 8C illustrates the magnetic flux density waveform Y (of magnetic core 12), with the vertical axis scaled linearly from −100% (negative saturation) to +100% (positive saturation). Ideal burden compensation is assumed, so the flux density remains very close to zero.
FIG. 8D illustrates the waveform of the secondary electric current error J2E. Since secondary current error J2E is closely related to the magnetic flux density Y of the magnetic core, and the flux density remains near zero as shown in FIG. 8C, current error J2E is near zero.
FIG. 8E illustrates the waveform of secondary electric current J2. Since secondary current error J2E is near zero, the waveform of secondary current J2 is almost identical to primary current J1 (though secondary current J2 is smaller than primary current J1 by the turns ratio of current transformer CT).
FIGS. 9A to 9E illustrate the magnetic operation of a prior-art current transformer with electronic assist using a switched-voltage output (this is functionally similar to the configuration of FIGS. 5 and 6 if control circuit 3 operates to compensate for secondary burden).
FIG. 9A illustrates one possible primary current waveform for primary current J1. This is the same reference waveform as shown in FIGS. 7A and 8A.
FIG. 9B illustrates that voltage V1 is now controlled with a pulse-width-modulated type of control to compensate for secondary burden. Control circuit 3 controls voltage V1 so that its average value over a brief period of time is approximately equal to the continuous voltage shown in FIG. 8B. The pulse frequency shown was chosen for illustrative purposes; it is normally advantageous to have a pulse frequency that is fast relative to the highest frequency that is to be measured. The relatively slow pulse frequency shown exemplifies a small time lag that is inherent in a PWM control response; this is most visible immediately after time T3.
FIG. 9C illustrates the magnetic flux density waveform Y (of magnetic core 12), with the vertical axis scaled linearly from −100% (negative saturation) to +100% (positive saturation). The pulse-width-modulated voltage output provides burden compensation that is somewhat less than ideal, which results in a somewhat jagged flux waveform. However, flux still remains relatively close to zero, and flux control is still good.
FIG. 9D illustrates the waveform of the secondary electric current error J2E. Since secondary current error J2E is closely related to the magnetic flux density Y of the magnetic core (and the magnetic core does not saturate), current error J2E has a waveform that is similar to the waveform of magnetic flux Y. The current error waveform J2E is seen to be very small relative to the waveform of FIG. 7D (the case of the uncompensated current transformer).
FIG. 9E illustrates the waveform of secondary electric current J2. Since secondary current error J2E is very small, the waveform of secondary current J2 is almost identical to primary current J1.
FIGS. 9A to 9E illustrate how switched-voltage pulses may be used to approximate the burden-reducing compensation described in the 517 patent. Using similar principles, switched voltages can be utilized to approximate other operating modes described in the 517 patent:                (a) The induction level of a magnetic body may be caused to transition from a known induction level to a preferred induction level. (A preferred induction level of zero may be chosen to demagnetize a magnetic body).        (b) When the induction level is not known, a preferred induction level may be established by changing the induction level of the magnetic body from an unknown induction level to a known induction level (such as saturation) and then to the preferred induction level.        (c) A preferred induction level may be maintained by causing the induced voltage across the winding to have an average value near zero (or by causing the integral of induced voltage to not exceed a predetermined value).        (d) The induction level may be made to vary with time in a preferred manner, including matching a control signal that is proportional to a reference induction level.        
Refer to the 157 patent for additional information about utilizing switched voltages for magnetic control of current transformers.